1. Field
Embodiments of the present invention relate generally to the field of feedback amplifiers.
2. Background
The fully differential amplifier (FDA) has become a key component in signal processing applications where high speed and signal fidelity are essential system requirements. Advantages offered by fully differential amplifiers over single ended amplifiers include inherent resistance to external noise sources, reduced even order harmonics, and increased dynamic range. FIG. 1 shows a circuit 100 illustrating a general purpose FDA 110 in a typical data acquisition application. Within the scope of this discussion, the term “general purpose” will refer to an amplifier circuit generally having the following properties: 1) a pair of differential inputs, νip and νim; 2) a pair of differential outputs, νop and νom; 3) a common mode reference input, νcm, to control the common mode voltage level at the FDA's outputs; 4) a feedback network external to the chip (i.e., user configurable gain); 5) the ability to DC couple both inputs and outputs; and 6) the ability to drive the closed loop FDA from either a differential or single ended source. The FDA in this example shall be considered ideal, meaning infinite differential open loop gain and input impedance across the frequency domain. The closed loop gain of circuit 100, assuming an ideal FDA 110, is given by:
                                          v                          o              ,              dm                                =                                    2                                                f                  1                                +                                  f                  2                                                      *                          [                                                                    v                    sp                                    ⁡                                      (                                          1                      -                                              f                        1                                                              )                                                  -                                                      v                    sm                                    ⁡                                      (                                          1                      -                                              f                        2                                                              )                                                              ]                                      ,                            (        1        )            where νo,dm, the differential mode output voltage, is defined asνo,dm≡νop−νom,  (2)f1 and f2 are the respective gain factors for positive and negative feedback networks
                                          f            x                    =                                    R              gx                                                      R                fx                            +                              R                gx                                                    ,                            (        3        )            and νsp and νsm refer to the positive and negative terminals of the source. For the balanced feedback network case, Rf1=Rf2=Rf and Rg1=Rg2=Rg, and Equation 1 reduces to:
                                          v                          o              ,              dm                                =                                    v                              s                ,                dm                                      *                          (                                                R                  f                                                  R                  g                                            )                                      ,                                  ⁢        where                            (        4        )                                          v                      s            ,            dm                          ≡                              v            sp                    -                                    v              sm                        .                                              (        5        )            The output common mode voltage level, defined as
                                          v                          o              ,              cm                                ≡                                    (                                                v                  op                                +                                  v                  om                                            )                        2                          ,                            (        6        )            is controlled by means of a separate integrated amplifier circuit. The inverting terminal of this amplifier is connected to both output terminals of the FDA through a matched resistive divider network while the non-inverting terminal is connected to a reference voltage, Vcm. The closed loop gain of this amplifier, usually referred to as the common mode feedback amplifier (CMFB), is set to unity in most FDA integrated circuits.
FIG. 2 illustrates one possible implementation of the FDA 110 of FIG. 1. The architecture in FIG. 2 is based on a voltage feedback approach. Using this architecture, the balanced FDA gain relationship given in Equation 4 becomes:
                              v                      o            ,            dm                          =                              v                          s              ,              dm                                *                                                    (                                                      R                    f                                                        R                    g                                                  )                            [                              1                                  1                  +                                      1                                          2                      ⁢                                                                        G                          dm                                                ⁡                                                  (                          s                          )                                                                    ⁢                                                                        Z                          t                                                ⁡                                                  (                          s                          )                                                                    ⁢                      f                                                                                  ]                        .                                              (        9        )            This is the conventional closed loop gain expression for a linear voltage feedback FDA with finite open loop gain, where Gdm(s) is the transconductance of the differential input stage 210, and Zt(s) is the transimpedance gain of each half of the forward amplifier's signal path.
The amplifier's loop gain is given by:T(s)=2Gdm(s)Zt(s)f.  (10)This equation reveals the fundamental limitation of voltage feedback amplifier architectures, whether differential or single ended out, which is the direct proportionality of loop gain to the feedback factor f. As closed loop gain is increased (i.e., f is decreased), the loop gain is reduced, resulting in a corresponding reduction in closed loop bandwidth.
Another limitation of the voltage feedback FDA architecture depicted in FIG. 2 is the sensitivity of its high impedance inputs to parasitic capacitance at the νip and νim nodes. Analysis of the circuit of FIG. 2, introducing capacitors Cip and Cim at νip and νim respectively and ground, shows that the loop gain equation is modified by an additional high frequency pole:
                                          T            ⁡                          (              s              )                                =                                    2              ⁢                                                G                  dm                                ⁡                                  (                  s                  )                                            ⁢                                                Z                  t                                ⁡                                  (                  s                  )                                            ⁢              f                                      (                              1                +                                                      sC                    i                                    ⁢                                      R                    eq                                                              )                                      ,                            (        11        )            where it is assumed for simplicity that Cip=Cim=Ci and Req is simply Rf∥Rg. Typically, the parasitic capacitance contributions from PCB, package, and ESD protection circuits can add up to 1-2 pF. For example, a FDA configured for a gain of +1, with Rf equal to 500 ohms, will exhibit a parasitic pole between 318 MHz and 636 MHz. Any attempt to improve bandwidth by decompensation of the open loop amplifier will be limited by the loss of phase margin in the vicinity of this pole frequency.
The noise power spectral density at the differential output of the FDA is given by the following equation, where contributions from the resistive feedback network elements have been ignored for simplicity.
                              Pn          o                =                                                                                                                        (                                              2                        ⁢                                                  Vn                          i                                                                    )                                        2                                    +                                                            (                                              2                        ⁢                                                  In                          ip                                                ⁢                                                  R                                                      eq                            ⁢                                                                                                                  ⁢                            1                                                                                              )                                        2                                    +                                                                                                                                                (                                              2                        ⁢                                                  In                          im                                                ⁢                                                  R                                                      eq                            ⁢                                                                                                                  ⁢                            2                                                                                              )                                        2                                    +                                                            (                                              2                        ⁢                                                                              Vn                            cm                                                    ⁡                                                      (                                                                                          f                                1                                                            -                                                              f                                2                                                                                      )                                                                                              )                                        2                                                                                                          (                                                f                  1                                +                                  f                  2                                            )                        2                                              (        7        )            
Vni is the input referred voltage noise, Inip and Inim, are the input referred current sources at the positive and negative input terminals respectively, and Vncm is the input referred noise of the CMFB amplifier. Reqx is defined as Rfx∥Rgx. If the feedback networks are balanced, the noise equation reduces to:
                              Pn          o                =                                                                              (                                      2                    ⁢                                          Vn                      i                                                        )                                2                            +                                                                    (                                          2                      ⁢                                              R                        eq                                                              )                                    2                                *                                  (                                                            In                      ip                      2                                        +                                          In                      im                      2                                                        )                                                                                    (                                  2                  ⁢                  f                                )                            2                                .                                    (        8        )            Since the differential input to the voltage feedback FDA is high impedance, the noise contribution from circuit elements referred to the input as voltage sources will usually dominate the differential noise spectral density at the FDA output. The noise gain of the voltage feedback FDA will be proportional to the closed loop amplifier's feedback factor f.
      Pn    o    =            Vn      i      2              f      2      Thus, operation at higher signal path gains will necessarily be accompanied by higher noise levels at the FDA outputs.
In addition to voltage feedback, current feedback techniques have been applied in the design of high speed single ended output op-amps for a number of years. Advantages of current feedback (CFB) op-amps over their voltage feedback (VFB) counterparts include gain-bandwidth independence, generally higher slew rates, and improved even order non-linearity rejection. These performance benefits usually come at the price of less precision, smaller input common mode voltage range, and higher inverting input referred noise current. CFB techniques have not easily migrated to FDAs, primarily due to the difficulty in developing practical, low input impedance, fully differential input stage architectures with good common mode voltage rejection. However, the case for integrating current feedback techniques into the design of fully differential amplifiers is becoming more and more compelling as the demands of data acquisition systems continue to drive signal path components to provide high signal fidelity across much wider system bandwidths.
Previous attempts at designing a current feedback FDA have involved using two single-ended CFB amplifiers to form a differential amplifier. For example, U.S. Pat. No. 6,636,116, entitled “FULLY DIFFERENTIAL CURRENT FEEDBACK AMPLIFIER” and assigned to Texas Instruments Incorporated, discloses a CFB FDA that develops a differential low impedance input from the outputs of a pair of buffers, both referenced to a potential at their respective high impedance inputs. However, a dual, “un-balanced” input approach such as this results in an input stage that cannot discriminate between differential and common mode signals. In order to correct the balance error that results from such an implementation, the interfering input common mode signal must be cancelled elsewhere in the circuit. This is achieved by means of an additional circuit that replicates and 180° phase shifts the output current from one input buffer and combines it with the output current from the other, and vice versa. Moreover, in single ended input to differential output applications, where the common mode voltage at the FDA inputs is “boot strapped” to the differential voltage at the FDA outputs, holding the input buffer inputs to a fixed reference voltage could cause heavy current loading on each of the input buffers, and therefore introduce a significant distortion in the input stages of these circuits.